Differential Amplifier Design

Note
The underlying theory about current mirrors etc is skipped but might be added at some point.

Intro to Differential Amplifier Design

Differential amplifiers amplify a voltage difference at the input and are fundamental circuits in electronics. They are used in operational amplifier circuits and when ever a differential signal need to be amplified. Differential signals do have the advantage that they are immune to crosstalk and noise as those influence have no impact on the difference of this voltage levels. see SRAM technology.

Differential Amp Resistors

fig1

 

The principle is that a rising V_{in1} will reduce the voltage at V_{out1}. This will reduce the voltage over Q2 and will lead to a lower current in the right path which is causing V_{out2} to increase. We need a Resistor R_{SS} to make sure that both Transistors stay in Saturation Region.

 

Key Parameters
Gain

The Overall gain consists of 2 parts, the differential gain A_{vD} and the Common mode gain A_{vC}. Since we are interested to build up an differential amplifier, the common mode gain should be as small as possible.

The differential gain defines how much the voltage difference between V_{in1} and V_{in2} will be amplified whereas the common mode gain amplifies the input voltages itself (what we don't want to happen):

V_{out1}-V_{out2}=A_{vD}(V_{in1}-V_{in2})\pm A_{VC}\frac{V_{in1}+V_{in2}}{2}

Common Mode Amplification

We can calculate the amplification if we connect both input pins together. As both Transistors and Resistors now work in parallel they the resistance is half of the original value whereas the transistors become twice as strong which means that we can double the Width/Length ratio.

Common Mode Amplification

fig2

As you can see this is now a common Source stage.
The gain for a CS stage is
\left | A_{V}(CS) \right |= g_{m}R_{D} from which we can derive the formula for our amplification.

\left | A_{vC} \right |= \tfrac{\tfrac{R_{D}}{2}}{\tfrac{1}{2\sqrt{}2_{n}C_{ox}\tfrac{W}{L}I_{d}}+R_{SS}}

You can see from that equation that an infinite R_{SS} would lead to no common mode amplification as it would be an ideal current source.

Input Common Mode Range

As it had been discussed above the amplifier just works around an operating point. The range in which the amplifier works is defined as input common mode range (ICMR). We have linear dependency between the output voltages until one of the transistors leaves saturation. The INput common mode range is ~ 50% of V_{DD}-V{SS}

Offset

Transistors are not ideal because of several reasons during manufacturing therefore an 2 equal input values must not result in equal output voltages. We therefore add an Offset Voltage V_{OS} to compensate this problem. V_{OS} is usually in the range of 5-20mV.

V_{in1}-(V_{in2}+V_{OS})=V_{out1}-V_{out2}

Real amplifiers

In real live the difference of the input voltages is amplified and a voltage which is proportional to this difference is generated.

Differential Amplifier Symbol

fig3

Note that V_{out} is limited to V_{DD} as upper bound and V_{SS} as lower bound.

We also want to be able to amplify "negative" voltages, therefore we lower V_{SS} to a negative value. We can achieve the same effect by adding an offset to the input voltages. However, if we do so we generate a new reference voltage, analogue ground which is 0V and in the middle between V_{DD} and V_{SS}.

Making it single ended

We could change the circuit from fig1 in such a way that we just ignore one of the output voltage. The Problem with that is that both output voltages "move" in the opposite direction when the input voltage difference changes. By losing one of this voltages the gain would drop by 50%.

A better solution is to use a current mirror as pull up network.

Differential Amplifier Design

fig4

Lets say  V_{in1} is rising:
The current in  M1I_{D1}) is increasing while the current in M2I_{D2}) is decreasing by the same value.
But at the same time the voltage over M3 is decreasing which is pulling gate node of the current mirror to ground. This will cause the currents in M3 and M4 to get bigger which is also causing the output voltage to rise.

Also you can see that I have replaced R_{SS} by a NMOS. The reason is that resistors are difficult to manufacture on a chip. The bias voltage V_{b} determines the "resistance" of this pull down network.

Calculations
I_{SS}

The current in M5 is the sum of the currents that go through M1 and M2

I_{SS}=I_{D1}+I_{D2}

gain

The small signal gain may be obtained from

i_{out}=g_{md}v_{ID}

where g_{md} is the transconductance of the differential inputs.

v_{ID}=r_{out}i_{out}

where r_{out} is determined my the M2 and M4

r_{out}=\frac{1}{g_{ds2}+g_{ds4}}=\frac{1}{(\lambda_{ds2}+\lambda_{ds4})\frac{I_{SS}}{2}}

we obtain (explanation on factors might be added in the future)

A_{v}=\frac{v_{out}}{g_{ds2}+g_{ds4}}=\frac{2}{\lambda_{2}+\lambda_{4} }\sqrt{K'_{N}{\frac{W_{1}}{L_{1}}\frac{1}{I_{SS}}}}

Input Voltage Range

As mentioned earlier M1 and M2 need to be in saturation, therefore the input voltages need to be in a certain range.

Investigation:
sweep the common mode voltage (inputs connected) [will be simulated later]

We have to find the highest and lowest possible input voltage for which M1 and M2 are in saturation.

For a single transistor we know that we need to fulfill V_{ds}\geqslant V_{GS}-V_{T} to be in saturation region (and V_{GS}\geqslant V_{T} of course)

Maximum voltage

The maximum common mode voltage is the same as the gate voltage at M1. It is V_{DD} minus the voltage drop over Source to gate M3 plus the threshold voltage of M1.

V_{IC}(max)=V_{G1}(max)=V_{DD}-V_{SG3}-V_{DS1}+V_{GS1}=V_{DD}-V_{SG3}+V_{TN1}

Minimum voltage

M5 determines the minimum voltage as M5 also has to stay in saturation.

V_{DS5}=V_{IC}(min)-V_{SS}-V_{GS1}\Rightarrow V_{DS5}(sat)+V_{SS}+V_{GS1}\geqslant V_{IC}(min)

Slew Rate

The Slew rate determines how fast the output can charge or discharge a capacitive load C_{L} at the output V_{out}. All internal parasitic contribute of course also to the Slew rate. However, M5 defines this attribute

SR={\frac{I_{5}}{C_{L}}}

3dB-frequency

We dither a constant input voltage (which is inside the ICMR) with a AC voltage. The 3dB frequency is the frequency of the AC voltage at which the output signal of this AC is only 70% percent. It is determined by the output resistance.

\omega_{-3dB}=2\pi f_{-3dB}=\frac{1}{R_{out}C_{L}}

Power dissipation

The power consumption is the voltage drop over the full circuit times the current in M5

P_{diss}=(V_{DD}-V_{SS})I_{SS}

 

Example Design

This example will be based on a 0.8µm Technology for which we have

 Parameter Description NMOS PMOS Unit
 V_{th0} Threshold Voltage  0,5\pm 0,05  -0,65\pm 0,05  V
 K' Transistor-Transconductance   175\pm 10%  60\pm 10%  \frac{\mu A}{V^{2}}
 \gamma Bodyfactor  0,58 0,42  V^{\frac{1}{2}}
 \lambda Channel-length-modulation factor 0,06@L=1 \mu m
0,04@L=2 \mu m
 0,06@L=1 \mu m
0,04@L=2 \mu m
 V^{-1}
 2 \left | \phi_{F} \right | Surface potential for strong
inversion
 0,8  0,8  V
Design Targets

Supply Voltage:
V_{DD}=3,3V
V_{SS}=0V

Slew Rate:
SR\geq20\frac{V}{\mu s}

3dB frequency
f_{-3dB}\geq200kHz(C_{L})=5pF

Input common mode range
0,75V\geq ICMR\geq 2,95V

Differential Amplification
A_{diff}=200

Power Dissipation
P_{diss}\geq 1mW

Calculation
I_{5}

The Slew Rate tells us how fast the load need to be charged/uncharged. As C_{L} is given (5pF) we can calculate the minimum current I_{5} (note that the Slew rate is given as Volts per µs and therefore needs to be multiplied)

I_{5}\geq SR*C_{L}= (20*10^{6}V)*(5*10^{-12}F)=100\mu A

The Power dissipation P_{diss} is giving us the upper bound for I_{5}

I_{5}=\leq \frac {P_{diss}}{V_{DD}-V{SS}} = \frac {1+10^{-3}W}{3,3V}=303\mu A

If we also take the channel length modulation factors into consideration we have

r_{out}=\frac{1}{g_{ds2}+g_{ds4}}=\frac{1}{(\lambda_{ds2}+\lambda_{ds4})\frac{I_{SS}}{2}}\Rightarrow I_{5}=\frac {2}{(\lambda_{ds2}+\lambda_{ds4})R_{out}}

And by having knowledge about the 3dB frequency we can calculate R_{out}

\omega_{-3dB}=2\pi f_{-3dB}=\frac{1}{R_{out}C_{L}}

\therefore 2\pi 200kHz \leq \frac{1}{R_{out}5pF}\Rightarrow R_{out} \leq \frac{1}{2\pi 200kHz*5pF}=159k\Omega

With knowledge about  R_{out} we can set a lower bound which is considering the channel length modulation factor.

I_{5}=\frac {2}{(\lambda_{ds2}+\lambda_{ds4})R_{out}} = \frac {2}{(0,06+0,06)V^{-1}*159k\Omega}\approx 105\mu A

We now know 100\mu A\leq105\mu A\leq I_{5}\leq 303\mu A

To have some margin we set I_{5}=200\mu A.

Designing the current mirror (ICMR max)

We need to determine the saturation current of M3 (and therefore also M4) to make sure that we have half of I_{ss} aka I_{5} balanced on each side \frac{I_{5}}{2}=100\mu A

we know that

V_{IC}(max)=V_{G1}(max)=2,95V=V_{DD}-V_{SG3}-V_{DS1}+V_{GS1}=V_{DD}-V_{SG3}+V_{TN1}

therefore

V_{SG3}=V_{DD}-V_{IC}(max)+V_{TN1}=3,3V-2,95V+0,5V=0,85V

Now we can determine the size of M3 and M4, the current is

I_{D3}=\frac{1}{2}K'\frac{W_{3}}{L_{3}}(V_{GS3}-V_{TP})^{2}

Note that this is an PMOS device and that V_{GS}=-V_{SG}

100\mu A=\frac{1}{2}60\frac{\mu A}{V^{2}}\frac{W_{3}}{L_{3}}(V_{GS3}-V_{TP})^{2}

100\mu A=\frac{1}{2}60\frac{\mu A}{V^{2}}\frac{W_{3}}{L_{3}}(-0,85+0,65)^{2}

\frac{W_{3}}{L_{3}}=\frac{100\mu A}{\frac{1}{2}60\frac{\mu A}{V^{2}}(-0,85+0,65)^{2}}=\frac{W_{4}}{L_{4}}= 83,33

Designing the Pull down transistors M1 and M2

From the gain specification and from the knowledge about the currents we now can calculate the W/L ratio for M1 and M2.

A_{diff}=g_{m1}R_{out}=\frac{g_{md}}{g_{ds2}+g_{ds4}}=\frac{\sqrt{2\mu_{n}C_{ox}\frac{W_{1}}{L_{1}}}}{(\lambda_{n}+\lambda_{p})\sqrt{I_{ss}}}

\therefore \frac{W_{1}}{L_{1}}=\frac{(\frac{A_{v}(\lambda_{n}+\lambda_{p})\sqrt{I_{5}}}{2})^{2}}{K'}=\frac{(\frac{200(0,06+0,06)\sqrt{200\mu A}}{2})^{2}}{175\frac{\mu A}{V^{2}}}\approx 164,57

M5

From the ICMR we can calculate V_{DS5}

V_{DS5}(sat)+V_{SS}+V_{GS2}\geq V_{IC}(min)
\therefore V_{DS5}(sat)= V_{IC}(min)-V_{SS}-V_{GS2}

We first need to know V_{GS2}

I_{2}=\frac{\mu_{n}C_{ox}}{2}\frac{W_{2}}{L_{2}}(V_{GS2}-V_{TN})^{2}

\therefore V_{GS2}=\sqrt{\frac{100\mu A}{\frac{175\frac{\mu A}{V^{2}}}{2}164,5}}=0,583V
\therefore V_{ds5}(sat)=0,166

W/L ratio for M_{5}

I_{5}=\frac{\mu_{n}C_{ox}}{2}\frac{W_{5}}{L_{5}}(V_{DS5}(sat))^{2}

\therefore \frac{W_{5}}{L_{5}}=\frac{I_{5}}{\frac{\mu_{n}C_{ox}}{2}(V_{ds5}(sat))^{2}}=\frac{200\mu A}{\frac{175 \frac{\mu A}{V^{2}}}{2}(0,166V)^{2}}\approx 82.29

V_{gs} for bias current mirror

I_{6}=I_{5}=\frac{\mu_{n}C_{ox}}{2}\frac{W_{5}}{L_{5}}(V_{GS6}-V_{TN})^{2}

V_{GS6}=\sqrt{\frac{200\mu A}{\frac{175\frac{\mu A}{V^{2}}}{2}82,29}}+0,5=0,666V

R=\frac{V_{DD}-V_{gs6}-V_{ss}}{I_{d6}}=\frac{3,3V-0,66V-0V}{200\mu A}=13,166k \Omega

 

Simulation

I use LTSPICE for the simulation.

Offset and transfer

First I want to check if the offset voltage and the transfer characteristics.
We set the negative input to analog 0V and sweep the positive input.

01_off_trans

Plot 1 V(3)

From the plot you can see that the supply voltage range is used from 0,9V to 3,1V
We do have an offset but as we are in the linear region this can be fixed by adding an offset voltage of about 3,5mV to the negative input.

It also can be seen that a 0,001V change of the input voltage results in a 0,2V difference at the output, so we have met our design goal for an Amplification factor of 200.

ICMR with unity gain configuration

Not we want to check the range of operation if we connect the output to the negative input and sweep the positive input.

plot 2 ICMR

We can see that we have met our design goal of 0,75V\geq ICMR\geq 2,95V

Bode Plot

In this configuration we want to see at which frequency the AC is not amplified anymore.

Plot 3 Bode Diagram

Plot 3 Bode Diagram

We have a -3dB frequency of around 355kHz which is better that the requirement of 200kHz so we also met this criteria.

Slew Rate

How fast can the capacitive load be charged and uncharged?

04_SR

We can see that the leading and the trailing edge need ~30ns to reach the desired voltage, the slew rate therefore is (the 10^{6} is needed to bring it to the \frac{V}{\mu s} scale

SR=\frac {1V}{30*10^{-9}V*10^{6}}=33.33\frac{V}{\mu s}\geq 20\frac{V}{\mu s}

so also the slew rate is better than our requirement.

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